High-speed switching devices such as bipolar transistors, MOSFETs and IGBT's enable increased carrier frequency for voltage-source PWM inverters, thus leading to much better operating characteristics. High-speed switching, however, causes the following serious problems, originating from a high rate-of-change in voltage and/or current:                a) ground current escaping to earth through stray capacitors inside motors and through long cables;        b) conducted and radiated EMI;        c) motor bearing current and shaft voltage; and        d) shortening of insulation life of motors and transformers.        
The voltage and/or current change caused by high-speed switching produces high-frequency oscillatory common-mode and normal-mode currents when the switching device(s) change state because parasitic stray capacitance inevitably exists inside a load, for example, an ac motor, as well as inside the switching converter. Thus, each time an inverter switching event occurs, the potential of the corresponding inverter output terminal moves rapidly with respect to ground, and a pulse of common mode current flows in the d-c link to the inverter, via the capacitance of the heatsink motor cable and motor windings to ground. The amplitude of this pulse of current for a class B (residential) motor drive is typically several hundred millamps to several amps; and the pulse width is typically 250 to 500 ns. For a class A drive (Industrial), and depending on the size of the motor and length of the motor cable, the pulse current amplitude is typically several amperes with a pulse width of 250 ns to 500 ns, to 20 amperes or more with a pulse width of 1 to 2 micro seconds.
The common mode oscillatory currents may have a frequency spectrum range from the switching frequency of the converter to several tens of MHZ, which creates a magnetic field and will produce radiated electromagnetic interference (EMI) throughout, thus adversely affecting electronic devices such as radio receivers, medical equipment, etc.
A number of Governmental restrictions apply to the degree of permissible line current EMI and permissible ground current in certain motor applications. Thus, in Class B residential (appliances), applications, ground current must be kept below from 1 to 20 mA over a frequency range from 0 to 30 kHz respectively (over a logarithmic curve); and conducted line current EMI must be kept below designated values (less than about 60 dBμV) over a frequency range of 150 kHz to 300 MHZ. For motor drive applications designated as class A Industrial applications, limitations on ground current are less stringent, but line current EMI is still limited over the 150 kHz to 30 MHZ range.
Generally, common-mode chokes and EMI filters, based on passive elements, may not completely solve these problems. Passive filters, consisting of a common mode inductor and “Y” capacitors in the input ac line have been used to filter the common mode current in such motor drive circuits. Passive common mode filters may place limits on the PWM frequency which can be used, are physically large (frequently a major fraction of the volume of the motor drive structure) and are expensive. Further, they are functionally imperfect in that they exhibit undesired resonance which runs counter to the desired filtering action. Further, in general purpose industrial drives, the drive circuit and motor are often connected by cables which are up to 100 meters or more long. The longer the cable, the greater the conducted common mode EMI in the motor cable, and the larger the required size of a conventional passive common mode input filter.
A common-mode transformer with an additional winding shorted by a resistor is known which can damp the oscillatory ground current. Unfortunately, a small amount of aperiodic ground current will still remain in this circuit.
Active filters for control of the common mode current in a pulse width modulated (PWM) controlled motor drive circuit are well known. Such devices are typically described in the paper an Active Circuit for Cancellation of Common-Mode Voltage Generated by a PWM Inverter, by Satoshi Ogasawara et al., IEES Transactions on Power Electronics, Vol. 13, No. 5, (September 1998 and in U.S. Pat. No. 5,831,842 in the names of Ogasawara et al.
FIG. 1 shows a typical prior art active filter circuit or EMI and noise canceller for an a-c motor. Thus, in FIG. 1, an a-c source comprising an input terminal L and a neutral terminal are connected to the a-c input terminals of a full wave bridge connected rectifier 40. While a single phase supply is shown, the principles in this and in all Figures to be described can be carried out with a three-phase or multi-phase input. The positive and negative busses of rectifier 40 contain points A and D respectively and are connected to a three-phase bridge connected PWM controlled inverter 41, at inverter terminals B and F. The output a-c terminals of the inverter are connected to a-c motor 42. A filter capacitor 40a is also connected across terminals B and F. Motor 42 has a grounded housing connected to ground wire 43 with ground terminal 43a. 
The active filter consists of a pair of transistors Q1 and Q2, connected across the d-c output lines of rectifier 40 with their emitters connected at node E. These define amplifiers which are controlled by output winding 44 of a differential transformer having input windings 45 and 46 connected in the positive and negative output busses of rectifier 40. The winding polarities are designated by the conventional dot symbols. Winding 44 is connected between the control terminals of transistors Q1 and Q2 and the common emitter node E. A d-c isolating capacitor 47 is connected to ground line 43 at node C.
The active filter including capacitor 47 defines a path for diverting the majority of the common mode current which can otherwise flow in the path L or N, A, B, M (motor 42), 43, 43a and back to L or N; (or in the reverse path when polarity reverses) or in path L or N, D, F, M, 43, 43a (or in the reverse path when polarity reverses). Thus, most common mode current can be diverted, for currents from positive terminal A, in the path B, M, C, E, Q2, F, B, for “positive current”, and in the pattern B, M, C, E, Q1, B for “negative” current by the proper control of transistor Q1, and Q2. The path for common mode current flowing into negative terminal D follows the path F, M, C, E, Q2, F for “positive” current and F, M, C, E, Q1, B for “negative” current. The degree of diversion depends on the current gain of winding 44 and the current gain of Q2, for “positive current”, and the current gain of winding 44 and current gain of Q1, for “negative” current. In order to obtain a sufficient degree of diversion of the common mode current, the overall current gain of winding 44 and transistors Q1 and Q2 must be high.
The sensing transformer 44, 45, 46 of FIG. 1 has been large and expensive in order to provide sufficiently high current gain. It would be very desirable to reduce the size and cost of this transformer without jeopardizing the operation of the circuit. A further problem is that because of the high gain required, this closed-loop circuit has a tendency to produce unwanted oscillation.
Further, it has been found that the transistors Q1 and Q2 may not be able to operate in their linear regions over a large enough range within the “headroom” defined by the circuit, thus defeating the active filtering action. The headroom, or the voltage between the collector and emitter of transistors Q1 and Q2 is best understood by considering the approximate equivalent circuit of FIG. 1, as shown in FIG. 2, in which the ground potential at C is the same as that of the neutral line in FIG. 1. Transistors Q1 and Q2 are shown as resistors R1 and R2 respectively with respective parallel connected diodes. The d-c bridge 40 is shown as two d-c sources 50 and 51, each producing an output voltage of VDC/2 where VDC is the full output voltage between the positive and negative busses at terminals A and D, and an a-c source 52 having a peak a-c voltage of VDC/2.
It can be seen from FIG. 2 that headroom can disappear at different portions of the cycle of source 52. Thus, consider a first situation in which the leakage impedances of transistors Q1 and Q2 are the same. In this case, the values of resistors R1 and R2 in FIG. 2 are about equal. Now, as the ground potential at terminal C swings between (+)VDC/2 and (−)VDC/2 with respect to the d-c midpoint at node 53 in FIG. 2, the potential at the emitters of transistors Q1 and Q2 also swings between (+)VDC/2and (−) VDC12, if it is assumed that the impedance of capacitor 47 is much smaller than R1 and R2. Therefore, during the periods when the potential at node E is close or equal to the potential of the d-c bus (at points B or F), insufficient voltage headroom exists for the relevant transistors Q1 or Q2 to operate as linear amplifiers, and the active filtering action is lost.
Consider next a condition in which the ground potential at C is the same as that of the neutral input line N in FIG. 1, and the leakage current of transistor Q2 is now much higher than that of transistor Q1, which, in FIG. 2 would be represented by the condition that R2 is much less than R1. This then biases the potential at E toward the negative d-c bus (at F). Therefore, the potential at E resides at the negative bus potential for a significant portion of each input cycle. During this period, transistor Q2 cannot operate as a linear amplifier and the active filtering action is lost.
Consider next the condition where the ground is at N and the resistance of transistor Q1 is much less than that of transistor Q2; that is, R1 is much less than R2. This would bias point E toward the positive d-c bus (point B) so that, for a significant portion of each input cycle, transistor Q1 cannot operate as a linear amplifier.
Lastly, consider a condition in which a small a-c potential (dotted line a-c source 52a in FIG. 2) exists between the ground wire 43 and neutral N of FIG. 1, as would occur if the grounded neutral of the supply transformer is electrically remote from the ground connection of the motor drive itself. In that case, the ground potential will swing between +(dV+VDC2/) and −(dV+VDC/2) where dV is the peak of the voltage wave shape of source 52a in FIG. 2. If the leakage characteristics of transistors Q1 and Q2 are about equal, the potential at E will attempt to swing by dV above the positive bus and by (−) dV below the negative d-c bus. During these periods, the voltage at E is clamped to the bus voltage by the low impedance reverse characteristics of transistors Q1 and Q2. Thus, no voltage headroom exists for the transistors to operate during these periods.
It would be very desirable to provide a circuit which provides sufficient headroom for transistors Q1 and Q2 under the above described conditions and for clamping the headroom voltages to a prescribed minimum level; or for regulating the average voltage at point E to the d-c midpoint potential.
The active filter of the prior art, as shown in FIG. 1, is always connected across the full d-c bus voltage. This requires a high enough voltage rating for the transistors Q1 and Q2 and causes a relatively high power dissipation in the active filter components. It would therefore be desirable to operate the active filter at a lower voltage, if possible, without degrading the performance of the active filter.